Phase-Locked Loops for High-Frequency Receivers and Transmitters - Part 3

The first part of this series introduced phase-locked loops (PLLs), described basic architectures and principles of operation. It also included an example of where a PLL is used in communications systems. In the second part of the series, critical performance specifications, like phase noise, reference spurs, and output leakage, were examined in detail, and their effects on system performance were considered. In this, the last part of the series, we will deal with some of the main building blocks that go to make up the PLL synthesizer. We will also compare integer-N and fractional-N architectures. The series will end with a summary of VCOs currently available on the market and a listing of the Analog Devices family of synthesizers.

PLL Synthesizer basic building blocks

A PLL synthesizer can be considered in terms of several basic building blocks. Already touched upon, they will now be dealt with in greater detail:

Phase-frequency detector (PFD)
Reference counter (R)
Feedback counter (N)

The Phase-Frequency Detector (PFD)

The heart of a synthesizer is the phase detector—or phase-frequency detector. This is where the reference frequency signal is compared with the signal fed back from the VCO output, and the resulting error signal is used to drive the loop filter and VCO. In a digital PLL (DPLL) the phase detector or phase-frequency detector is a logical element. The three most common implementations are:

Exclusive-or (EXOR) gate
J-K flip-flop
Digital phase-frequency detector

Here we will consider only the PFD, the element used in the ADF4110 and ADF4210 synthesizer families, because—unlike the EXOR gate and the J-K flip flop—its output is proportional to the frequency difference between the two inputs when it is in the unlocked state.

Figure 1 shows one implementation of a PFD, basically consisting of two D-type flip flops. One Q output enables a positive current source; and the other Q output enables a negative current source. Assuming that, in this design, the D-type flip flop is positive-edge triggered, the states are these:

11–both outputs high, is disabled by the AND gate (U3) back to the CLR pins on the flip flops.

00–both P1 and N1 are turned off and the output, OUT, is essentially in a high impedance state.

10–P1 is turned on, N1 is turned off, and the output is at V+.

01–P1 is turned off, N1 is turned on, and the output is at V–. 

Consider now how the circuit behaves if the system is out of lock and the frequency at +IN is much higher than the frequency at -IN, as exemplified in Figure 2.

Since the frequency at +IN is much higher than that at –IN, the output spends most of its time in the high state. The first rising edge on +IN sends the output high and this is maintained until the first rising edge occurs on –IN. In a practical system this means that the output, and thus the input to the VCO, is driven higher, resulting in an increase in frequency at –IN. This is exactly what is desired.

If the frequency on +IN were much lower than on –IN, the opposite effect would occur. The output at OUT would spend most of its time in the low condition. This would have the effect of driving the VCO in the negative direction and again bring the frequency at –IN much closer to that at +IN, to approach the locked condition. Figure 3 shows the waveforms when the inputs are frequency-locked and close to phase-lock.

Since +IN is leading –IN, the output is a series of positive current pulses. These pulses will tend to drive the VCO so that the –IN signal become phase-aligned with that on +IN.

When this occurs, if there were no delay element between U3 and the CLR inputs of U1 and U2, it would be possible for the output to be in high-impedance mode, producing neither positive nor negative current pulses. This would not be a good situation. The VCO would drift until a significant phase error developed and started producing either positive or negative current pulses once again. Over a relatively long period of time, the effect of this cycling would be for the output of the charge pump to be modulated by a signal that is a subharmonic of the PFD input reference frequency. Since this could be a low frequency signal, it would not be attenuated by the loop filter and would result in very significant spurs in the VCO output spectrum, a phenomenon known as the backlash effect. The delay element between the output of U3 and the CLR inputs of U1 and U2 ensures that it does not happen. With the delay element, even when the +IN and –IN are perfectly phase-aligned, there will still be a current pulse generated at the charge pump output. The duration of this delay is equal to the delay inserted at the output of U3 and is known as the anti-backlash pulse width.

The Reference Counter

In the classical Integer-N synthesizer, the resolution of the output frequency is determined by the reference frequency applied to the phase detector. So, for example, if 200-kHz spacing is required (as in GSM phones), then the reference frequency must be 200 kHz. However, getting a stable 200-kHz frequency source is not easy. A sensible approach is to take a good crystal-based high frequency source and divide it down. For example, the desired frequency spacing could be achieved by starting with a 10-MHz frequency reference and dividing it down by 50. This approach is shown in the diagram in Figure 4.

The Feedback Counter, N

The N counter, also known as the N divider, is the programmable element that sets the relationship between the input and output frequencies in the PLL. The complexity of the N counter has grown over the years. In addition to a straightforward N counter, it has evolved to include a prescaler, which can have a dual modulus.

This structure has grown as a solution to the problems inherent in using the basic divide-by-N structure to feed back to the phase detector when very high-frequency outputs are required. For example, let's assume that a 900-MHz output is required with 10-kHz spacing. A 10-MHz reference frequency might be used, with the R-Divider set at 1000. Then, the N-value in the feedback would need to be of the order of 90,000. This would mean at least a 17-bit counter capable of dealing with an input frequency of 900 MHz.

To handle this range, it makes sense to precede the programmable counter with a fixed counter element to bring the very high input frequency down to a range at which standard CMOS will operate. This counter, called a prescaler, is shown in Figure 5.

However, using a standard prescaler introduces other complications. The system resolution is now degraded (F₁ × P). This issue can be addressed by using a dual-modulus prescaler (Figure 6). It has the advantages of the standard prescaler but without any loss in system resolution. A dual-modulus prescaler is a counter whose division ratio can be switched from one value to another by an external control signal. By using the dual-modulus prescaler with an A and B counter one can still maintain output resolution of F₁. However, the following conditions must be met:

1. The output signals of both counters are High if the counters have not timed out.
2. When the B counter times out, its output goes Low, and it immediately loads both counters to their preset values.
3. The value loaded to the B counter must always be greater than that loaded to the A counter.

Assume that the B counter has just timed out and both counters have been reloaded with the values A and B. Let's find the number of VCO cycles necessary to get to the same state again.

As long as the A counter has not timed out, the prescaler is dividing down by P+1. So, both the A and B counters will count down by 1 every time the prescaler counts (P+1) VCO cycles. This means the A counter will time out after ((P+1) × A) VCO cycles. At this point the prescaler is switched to divide-by-P. It is also possible to say that at this time the B counter still has (B-A) cycles to go before it times out. How long will it take to do this: ((B-A) × P). The system is now back to the initial condition where we started.

The total number of VCO cycles needed for this to happen is:

When using a dual-modulus prescaler, it is important to consider the lowest and highest values of N. What we really want here is the range over which it is possible to change N in discrete integer steps. Consider the expression N = A + BP. To ensure a continuous integer spacing for N, A must be in the range 0 to (P–1). Then, every time B is incremented there is enough resolution to fill in the all the integer values between BP and (B+1)P. As was already noted for the dual-modulus prescaler, B must be greater than or equal to A for the dual modulus prescaler to work. From these we can say that the smallest division ratio possible while being able to increment in discrete integer steps is:

The highest value of N is given by

In this case A_max and B_max are simply determined by the size of the A and B counters.

Now for a practical example with the ADF4111.
Let's assume that the prescaler is programmed to 32/33.
A counter: 6 bits means A can be 2⁶ - 1 = 63
B counter: 13 bits means B can be 2¹³ - 1 = 8191

ADF4110 Family

The building blocks discussed in the previous sections are all used in the new families of integer-N synthesizers from ADI. The ADF4110 family of synthesizers consists of single devices and the ADF4210 family consists of dual versions. The block diagram for the ADF4110 is shown below. It contains the reference counter, the dual-modulus prescaler, the N counter and the PFD blocks described above.

Fractional-N Synthesizers

The authors are indebted to The McGraw-Hill Companies for permission to use copyrighted material from Reference 4 in this section.

Many of the emerging wireless communication systems have a need for faster switching and lower phase noise in the local oscillator (LO). Fractional-N synthesis is a means of achieving both of these aims.

The technique was originally developed in the early 1970s.This early work was done mainly by Hewlett Packard and Racal. The technique originally went by the name of "digiphase" but it later became popularly named fractional-N.

In the standard synthesizer, it is possible to divide the RF signal by an integer only. This necessitates the use of a relatively low reference frequency (determined by the system channel spacing) and results in a high value of N in the feedback. Both of these facts have a major influence on the system settling time and the system phase noise. The low reference frequency means a long settling time, and the high value of N means larger phase noise.

If division by a fraction could occur in the feedback, it would be possible to use a higher reference frequency and still achieve the channel spacing. This lower fractional number would also mean lower phase noise.

If fact it is possible to implement division by a fraction over a long period of time by alternately dividing by two integers (divide by 2.5 can be achieved by dividing successively by 2 and 3).

So, how does one divide by X or (X + 1) (assuming that the fractional number is between these two values)? Well, the fractional part of the number can be allowed to accumulate at the reference frequency rate.

The diagram of Figure 9 shows the timing of the Fractional-N system described in Figure 8. For the purpose of this example, we have assumed a divide ratio of 4.6.

The signal F_OUT shows 46 cycles during the time that FREF is executing 10 cycles. During the time that FREF generates its first cycle, the N counter is required to divide by 4.6. Of course, this is not possible. It divides by 4. Thus, in the first cycle, 0.6 pulses are "missing" from the counter output. This is memorized in the system using an accumulator. The accumulator uses the same code as the F Register. At the beginning of each reference cycle, the accumulator adds the F Register contents to its previously accumulated value. Thus, starting at time 0, the accumulator will keep track of the "missing" pulse fractions.

In the second reference cycle, the N counter will again divide by 4. The accumulator will now add 0.6 to the 0.6 accumulated from the first Reference Cycle. This gives a value of 1.2 in the accumulator but, since it can only store values less than 1, an overflow will be generated and 0.2 kept as the accumulator contents.

The overflow is used to activate the pulse removing circuit. Thus, the next pulse generated by the VCO is removed from the input to the N counter. This pulse removal has the same effect as dividing by 5 instead of 4. As shown in the diagram, the accumulator again overflows in the 4th, 5th, 7th, 9th and 10th cycles. So, in a series of 10 reference cycles there are six overflows yielding a total count of (4 × 4) + (6 × 5) = 46. This is exactly what was wanted.

Summary of VCO Manufacturers

With the explosive growth in wireless communications, the demand for products like synthesizers and VCOs has increased dramatically over the past five years. Interestingly, until now, the markets have been served by two distinct sets of manufacturers. Below is listed a selection of players in the VCO field. This list is not meant to be all-inclusive, but rather gives the reader a feel for some of the main players.

VCOs

Murata Murata has both 3-V and 5-V devices available. The VCOs are
mainly narrowband for wireless handsets and base stations.
Frequencies are determined by the wireless frequency standards.

Vari-L Vari-L addresses the same market as Murata. 3-V and 5-V
devices are available.

Alps Alps makes VCOs for wireless handsets and base stations.

Mini-Circuits Mini-Circuits offers both narrowband and wideband VCOs.

Z-Comm Z-Communications has both wideband and narrowband VCOs.
The wideband VCOs typically have an octave tuning range
(1 GHz to 2 GHz, for example) and operate from a supply voltage
of up to 20 V. They offer surface-mount packaging.

Micronetics Micronetics offers both narrowband and wideband VCOs.
Their strength lies more in the wideband products where they
can go from an octave range at anything up to 1200 MHz.
Above these output frequencies, the range is somewhat reduced.

The Analog Devices Synthesizer Family

The following table lists the members of the ADF4xxx synthesizer family. It includes single and dual, and integer-N and fractional-N devices. Currently available preliminary data sheets are indicated by links. For samples, availability, and further information, get in touch with linear.applications@analog.com.

ADI PLL Selection Guide - February 2002

References

1. Mini-Circuits Corporation, VCO Designers' Handbook, 1996.
2. L.W. Couch, Digital and Analog Communications Systems, Macmillan Publishing Company, New York, 1990.
3. P. Vizmuller, RF Design Guide, Artech House, 1995.
4. R.L. Best, Phase Locked Loops: Design, Simulation and Applications, 3rd edition, McGraw-Hill, 1997.
5. Goldberg, Bar-Giora, Digital Frequency Synthesis Demystified, Eagle Rock, VA: LLH Technology Publishing, 1999.

Acknowledgements

The ADF4xxx Family of synthesizers is designed at the Analog Devices facility in Limerick, Ireland. The product line team includes: Mike Tuthill, Leo McHugh, Bill Hunt, Mike Keaveney, Brendan Daly, Paul O'Brien, Paul Mallon, Ian Collins, Sinead O'Keefe, Liam McCann, Patrick Walsh, Cristoir O'Reilly, Paul Laven, Samuel Landete, Niall Kearney, and Mike Curtin. The group would like to acknowledge the valuable contributions of Jon Strange and Ashish Shah at Analog Devices, Kent (U.K.), and of Fred Weiss at Analog Devices Northwest Labs (Beaverton, OR).

Authors

Mike Curtin

Paul O'Brien